The present invention relates to ranging systems, including radars and acoustic navigation systems, which employ limiting constant false alarm rate receivers, and, in particular, to an improvement in such systems which reduces recognition jitter.
Acoustic ranging is used in ship navigation and in similar applications such as, for example, torpedo tracking.
A torpedo tracking system typically uses a one-way link. At known times, a transmitter in a torpedo initiates pulsed recognition signals at a fixed carrier-wave frequency. Each pulsed recognition signal is a tone-burst. The tone-bursts propagate through water to a plurality of acoustic pick-up transducers, or hydrophones, at known locations. The location of the torpedo is determined in a receiver by comparing the times of arrival of the tone-bursts at the several hydrophones.
A typical acoustic navigation system for ships uses a two-way link. On the ship, a ship set includes a transmitter, a receiver and an acoustic transducer mounted below the water line on the ship's hull. Transponders are at known locations on the ocean floor, each similarly including an acoustic transducer, a receiver and a transmitter. An interrogation signal pulse, or tone-burst, is initiated by the ship-set transmitter. When the interrogation tone-burst is recognized in the transponder receiver, the transponder transmitter initiates propagation of a reply tone-burst, which is subsequently recognized in the ship-set receiver. In order to avoid confusion between remotely transmitted signals and echo returns from a locally transmitted signal, the carrier-wave frequencies of the interrogation and reply signals are preferably selected to be different from each other. In addition, the carrier-wave frequencies of the transmitters of a plurality of transponders in the vicinity of each other may all be selected to be different from each other as a code to identify the source of a reply recognition signal.
Constant false alarm rate (CFAR) receivers have found broad application in acoustic ranging systems. They are also useful in radars. The characteristics of such receivers are discussed generally in Skolnick, Radar Handbook, New York, McGraw-Hill Book Co., 1970, Sec. 5.8, page 5-29 et seq. An implementation for limiting CFAR receivers is disclosed in U.S. Pat. No. 3,320,576, "Receiver For Processing A Plurality of Adjacent Closely Spaced Input Signals," issued May 16, 1977 to A. M. Dixon and Reginald J. Cyr. The latter co-inventor is the inventor herein. The entire disclosure of said patent is hereby incorporated by reference into this specification for its description of a CFAR receiver. Where the term CFAR receiver or constant false alarm rate receiver is used hereafter, a receiver of the type disclosed in said prior patent is intended to be indicated.
A CFAR receiver of the type disclosed in said prior patent uses a design in which the following functional elements typically operate in cascade: a broad-band input bandpass filter, a full limiter, one or more narrow-band bandpass filters, each of which operates to filter the limiter output, a detector for the output of each narrow-band bandpass filter, a smoothing filter for the output of each detector, and a threshold-responsive pulse-generating circuit for responding to the output of each smoothing filter.
The broad-band input bandpass filter minimizes the reception of noise at frequencies which are outside the range of all signal frequencies which it is desired to receive. The full limiter is a very high gain clipping amplifier which has a substantially constant amplitude output and therefore a substantially constant power output. Except for transitions between saturation limits, the limiter output is always saturated, even with no input to the receiver other than receiver self-noise or background noise from the environment. The frequency characteristics of the narrow-band bandpass filters are included within the passband of the broad-band filter. Each narrow-band filter characteristic is centered on the nominal carrier-wave frequency of a recognition signal which is desired to be received. This narrow-band frequency characteristic is made as narrow as is practical to reduce the probability of recognizing noise bursts as if they were signals when the noise power density is high at frequencies near the recognition signal carrier-wave frequency. In the marine environment, such noise bursts can emanate from, for example, ship engines, pumps and propellers. The selection of the bandwidth for a narrow-band bandpass filter is based on such considerations as, for example, pulse width of the recognition signal tone-burst, doppler frequency shift and drift of filter parameters due to such factors as, for example, variation in temperature and aging.
The bandwidth of the frequency characteristic for the broad-band input bandpass filter is typically selected to be about ten times the bandwidth of the narrow-band bandpass filter. This selection determines the bandwidth ratio for the CFAR receiver. The bandwidth ratio selected determines the extent to which noise which is passed by the broad-band input filter is suppressed at the output of the narrow-band filter in the absence of signal. Where the ratio of the bandwidth of the broad-band filter to the bandwidth of the narrow-band filter is ten-to-one, for example, the average noise power at the output of the narrow-band filter, in the absence of signal, is approximately one-tenth the average noise power at the output of the broad-band input filter.
In acoustic ranging systems, narrow-band bandpass filters typically have a bandwidth in the range from about 200 to about 600 hertz, while broad-band bandpass filters typically have a bandwidth in the range from about two to about six kilohertz.
Each smoothing filter integrates a detector output to provide a signal which is a measure of the energy in the passband of the associated narrow-band filter during a time interval of a predetermined length. The length of the smoothing time interval is determined by the smoothing filter parameters. When the smoothing filter output exceeds a preselected threshold value for recognition of signals, the associated threshold-responsive pulse-generating circuit generates a recognition pulse. The recognition pulse may be used, for example, to initiate reply signal transmission by a transponder transmitter, or, as another example, to determine elapsed time since the transmission of the interrogation signal in a range computer of a ship-set receiver in an acoustic ranging system.
When no signal is present at the input to a CFAR receiver, the limiter is controlled by a combination of self-noise and environmental noise. The limiter output waveform resembles a square wave. The amplitude of the waveform is fixed by the limiter clipping or saturation levels. However, the period of the waveform varies due to the noise. Under these same conditions, the narrow-band filter output waveform resembles a sine wave, but both the period and amplitude of the sine wave vary with time as nonlinear functions of the instantaneous noise characteristics. The smoothing filter output level fluctuates in response to the amplitude variations of the narrow-band filter output.
When a substantially constant-amplitude, monochromatic, pulsed signal, such as, for example, a tone-burst, is present at the input of a CFAR receiver, the noise-induced variations and fluctuations in the receiver are suppressed. The period of the waveform at the output of the limiter approaches the substantially constant, or monochromatic, period of the signal carrier wave. Similarly, at the output of that narrow-band filter the passband of which includes the signal carrier-wave frequency, the waveform approaches that of a sine wave of constant amplitude and period. The fluctuations in all smoothing filter output levels are reduced accordingly. The extent to which noise-induced variations and fluctuations are suppressed by the presence of signal in a CFAR receiver, is proportional to the signal-to-noise ratio.
Stated alternatively, as the signal present in the output of the limiter increases and acquires control over a portion of the substantially constant limiter output power, less of that power remains available for noise. Thus, the presence of signal suppresses the noise in a CFAR receiver including noise at the output of a smoothing filter, and, correspondingly, at the input to the associated threshold-responsive pulse-generating circuit.
For a sufficiently large signal-to-noise ratio, and given that the frequency of the signal is in the passband of a narrow-band filter, the associated smoothing filter output level rises above its average noise-only (absence of signal) level by a factor which is approximately equal to the bandwidth ratio. Increases in the signal-to-noise ratio beyond this point, however, produce no additional significant increase in the smoothing filter output level and corresponding threshold-responsive pulse-generating circuit input level. This is due to the limiting action of the limiter.
As has been indicated, recognition of an interrogation recognition signal pulse or a reply recognition signal pulse occurs in a CFAR receiver when the smoothing filter output and corresponding threshold-responsive pulse-generating circuit input exceeds a preselected recognition threshold value. The recognition threshold value selected determines the probability of signal recognition and the accompanying false alarm rate for a given signal-to-noise ratio. Once a tone-burst arrives at the input of the CFAR receiver, the mean time required for the smoothing filter output to rise to the recognition threshold value and trigger generation of a recognition pulse is a known or determinable function of system parameters and also a function of the average level of the smoothing filter output due to noise alone just prior to the time of arrival of the recognition signal tone-burst. These factors may be accounted for, and, if desired, corrected for in the calculation of range. However, another factor is not readily accounted for. The time-varying fluctuation in the level of the smoothing filter output about the average due to noise alone at the time a recognition signal tone-burst is available to increase the smoothing filter output level is a random variable. If this fluctuation has instantaneously raised the smoothing filter output level, less time will be required to drive that level up to the recognition threshold as the recognition signal arrives. If the fluctuation has instantaneously lowered the smoothing filter output level, more time will be required to drive that level up to the recognition threshold as the recognition signal arrives. The random error or uncertainty thus introduced into range calculations by noise is termed recognition time uncertainty or recognition jitter.
Since recognition jitter in a prior art ranging system using a CFAR receiver is primarily a function of the random noise variations occurring during the smoothing time interval just prior to the time a recognition signal is first sensed, it is substantially independent of signal-to-noise ratio. It is primarily a function of the noise alone. The error or uncertainty introduced into range calculations by recognition jitter does not decrease significantly as the signal-to-noise ratio increases beyond the minimum value required to raise the smoothing filter output above its noise-only level by a factor approximately equal to the bandwidth ratio of the receiver. This is a disadvantage of the prior art system. It is generally regarded as advantageous to design transmitter-receiver systems so that uncertainties due to noise decrease without limit as the signal-to-noise ratio increases and thereby improves.
Recognition jitter in a two-way link tends to be greater than in a one-way link. That is because of the uncertainty or error introduced into each of the two one-way links which are included in a two-way link. The recognition jitter for a two-way link is equal to the square root of the sum of the squares of the recognition jitter for the two included one-way links.
In currently used acoustic ranging systems, typical values for the carrier-wave frequency are in the range of from about seven kilohertz (7 KHz) to about sixteen kilohertz (16 KHz). Typical pulse durations for the recognition signal tone-bursts in these systems are in the range from about four milliseconds (4 ms) to about ten milliseconds (10 ms). For two-way links in such systems, the three-standard-deviation recognition jitter is typically about .+-.0.5 milliseconds.
Some prior art acoustic ranging systems have been designed in which the recognition jitter is reduced below that obtainable in the prior art systems using CFAR receivers discussed above. These improved systems employ digital-code correlation techniques. In these systems, a transmitter initiates transmission of a coded serial bit-stream, typically about thirty bits long, at a carrier frequency of about ten kilohertz. A receiver incorporates means for correlating a received bit-stream with a stored digital word having the intended predetermined thirty-bit code. If the correct code is received, the correlation function has a sharp peak which rises above the recognition threshold. The three-standard-deviation recognition jitter in such systems is typically about plus or minus one bit or about .+-.100 microseconds. This is an improvement of about five-to-one over the prior art ranging systems using CFAR receivers discussed above.
The relatively lower recognition jitter and accompanying higher accuracy range determination obtained in acoustic ranging systems using digital code correlation techniques requires a significant increase in equipment complexity and cost over that of acoustic ranging systems using CFAR receivers. In addition, the power required to be expanded is significantly increased in certain digital-code correlation technique embodiments. While increased power consumption may not be a serious deterrent to use of a given technique in a ship set, or a system having a short useful life as in the case of some torpedoes, increased power consumption is a strong negative factor in situations requiring a transponder placed on the ocean floor for long term use. Increased power consumption can be accommodated in such a transponder only by accepting a shorter life for the transponder's self-contained power supply, or by increasing the capacity of the power supply. It is desirable to avoid both of these alternatives.
Thus, there has long been a need for a reduced recognition jitter system that minimizes complexity and system power drain during its operational life.